Multiple-Output Dual-Polarity DC/DC Converters and Voltage Regulators

ABSTRACT

A multi-output dual polarity inductive boost converter includes an inductor, a first output node, a second output node, and a switching network, the switching network configured to provide the following modes of circuit operation: a first mode where the positive electrode of the inductor is connected to an input voltage and the negative electrode of the inductor is connected to ground; 2) a second mode the negative electrode of the inductor is connected to ground and the positive electrode of the inductor is connected in sequence to one or more of the fourth and fifth output nodes; and 3) a third mode where the positive electrode of the inductor is connected to the input voltage and the negative electrode of the inductor is connected in sequence to one or more of the first, second and third output nodes.

BACKGROUND OF THE INVENTION

Voltage regulation is commonly required to prevent variation in thesupply voltage powering various microelectronic components such asdigital ICs, semiconductor memory, display modules, hard disk drives, RFcircuitry, microprocessors, digital signal processors and analog ICs,especially in battery powered application likes cell phones, notebookcomputers and consumer products.

Since the battery or DC input voltage of a product often must bestepped-up to a higher DC voltage, or stepped-down to a lower DCvoltage, such regulators are referred to as DC-to-DC converters.Step-down converters are used whenever a battery's voltage is greaterthan the desired load voltage. Step-down converters may compriseinductive switching regulators, capacitive charge pumps, and linearregulators. Conversely, step-up converters, commonly referred to boostconverters, are needed whenever a battery's voltage is lower than thevoltage needed to power its load. Step-up converters may compriseinductive switching regulators or capacitive charge pumps.

Of the aforementioned voltage regulators, the inductive switchingconverter can achieve superior performance over the widest range ofcurrents, input voltages and output voltages. The fundamental principalof a DC/DC inductive switching converter is based on the simple premisethat the current in an inductor (coil or transformer) cannot be changedinstantly and that an inductor will produce an opposing voltage toresist any change in its current.

The basic principle of an inductor-based DC/DC switching converter is toswitch or “chop” a DC supply into pulses or bursts, and to filter thosebursts using a low-pass filter comprising and inductor and capacitor toproduce a well behaved time varying voltage, i.e. to change DC into AC.By using one or more transistors switching at a high frequency torepeatedly magnetize and de-magnetize an inductor, the inductor can beused to step-up or step-down the converter's input, producing an outputvoltage different from its input. After changing the AC voltage up ordown using magnetics, the output is then rectified back into DC, andfiltered to remove any ripple.

The transistors are typically implemented using MOSFETs with a lowon-state resistance, commonly referred to as “power MOSFETs”. Usingfeedback from the converter's output voltage to control the switchingconditions, a constant well-regulated output voltage can be maintaineddespite rapid changes in the converter's input voltage or its outputcurrent.

To remove any AC noise or ripple generated by switching action of thetransistors, an output capacitor is placed across the output of theswitching regulator circuit. Together the inductor and the outputcapacitor form a “low-pass” filter able to remove the majority of thetransistors' switching noise from reaching the load. The switchingfrequency, typically 1 MHz or more, must be “high” relative to theresonant frequency of the filter's “LC” tank. Averaged across multipleswitching cycles, the switched inductor behaves like a programmablecurrent source with a slow-changing average current.

Since the average inductor current is controlled by transistors that areeither biased as “on” or “off” switches, then power dissipation in thetransistors is theoretically small and high converter efficiencies, inthe eighty to ninety percent range, can be realized. Specifically when apower MOSFET is biased as an on-state switch using a “high” gate bias,it exhibits a linear I-V drain characteristic with a low R_(DS)(on)resistance typically 200 milliohms or less. At 0.5 A for example, such adevice will exhibit a maximum voltage drop I_(D)·R_(DS)(on) of only 100mV despite its high drain current. Its power dissipation during itson-state conduction time is I_(D) ²·R_(DS)(on). In the example given thepower dissipation during the transistor's conduction is (0.5A)²·(0.2Ω))=50 mW.

In its off state, a power MOSFET has its gate biased to its source, i.e.so that V_(GS)=0. Even with an applied drain voltage V_(DS) equal to aconverter's battery input voltage V_(batt), a power MOSFET's draincurrent I_(DSS) is very small, typically well below one microampere andmore generally nanoamperes. The current I_(DSS) primarily comprisesjunction leakage.

So a power MOSFET used as a switch in a DC/DC converter is efficientsince in its off condition it exhibits low currents at high voltages,and in its on state it exhibits high currents at a low voltage drop.Excepting switching transients, the I_(D)·V_(DS) product in the powerMOSFET remains small, and power dissipation in the switch remains low.

Power MOSFETs are not only used to convert AC into DC by chopping theinput supply, but may also be used to replace the rectifier diodesneeded to rectify the synthesized AC back into DC. Operation of a MOSFETas a rectifier often is accomplished by placing the MOSFET in parallelwith a Schottky diode and turning on the MOSFET whenever the diodeconducts, i.e. synchronous to the diode's conduction. In such anapplication, the MOSFET is therefore referred to as a synchronousrectifier.

Since the synchronous rectifier MOSFET can be sized to have a lowon-resistance and a lower voltage drop than the Schottky, conductioncurrent is diverted from the diode to the MOSFET channel and overallpower dissipation in the “rectifier” is reduced. Most power MOSFETsincludes a parasitic source-to-drain diode. In a switching regulator,the orientation of this intrinsic P-N diode must be the same polarity asthe Schottky diode, i.e. cathode to cathode, anode to anode. Since theparallel combination of this silicon P-N diode and the Schottky diodeonly carry current for brief intervals known as “break-before-make”before the synchronous rectifier MOSFET turns on, the average powerdissipation in the diodes is low and the Schottky oftentimes iseliminated altogether.

Assuming transistor switching events are relatively fast compared to theoscillating period, the power loss during switching can in circuitanalysis be considered negligible or alternatively treated as a fixedpower loss. Overall, then, the power lost in a low-voltage switchingregulator can be estimated by considering the conduction and gate drivelosses. At multi-megahertz switching frequencies, however, the switchingwaveform analysis becomes more significant and must be considered byanalyzing a device's drain voltage, drain current, and gate bias voltagedrive versus time.

Based on the above principles, present day inductor-based DC/DCswitching regulators are implemented using a wide range of circuits,inductors, and converter topologies. Broadly they are divided into twomajor types of topologies, non-isolated and isolated converters.

The most common isolated converters include the flyback and the forwardconverter, and require a transformer or coupled inductor. At higherpower, full bridge converters are also used. Isolated converters areable to step up or step down their input voltage by adjusting theprimary to secondary winding ratio of the transformer. Transformers withmultiple windings can produce multiple outputs simultaneously, includingvoltages both higher and lower than the input. The disadvantage oftransformers is they are large compared to single-winding inductors andsuffer from unwanted stray inductances.

Non-isolated power supplies include the step-down Buck converter, thestep-up boost converter, and the Buck-boost converter. Buck and boostconverters are especially efficient and compact in size, especiallyoperating in the megahertz frequency range where inductors 2.2 μH orless may be used. Such topologies produce a single regulated outputvoltage per coil, and require a dedicated control loop and separate PWMcontroller for each output to constantly adjust switch on-times toregulate voltage.

In portable and battery powered applications, synchronous rectificationis commonly employed to improve efficiency. A step-down Buck converteremploying synchronous rectification is known as a synchronous Buckregulator. A step-up boost converter employing synchronous rectificationis known as a synchronous boost converter.

Synchronous Boost Converter Operation: As illustrated in FIG. 1, priorart synchronous boost converter 1 includes a low-side power MOSFETswitch 4, battery connected inductor 5, an output capacitor 8, and“floating” synchronous rectifier MOSFET 7 with parallel rectifier diode6. The gates of the MOSFETs driven by break-before-make circuitry 3 andcontrolled by PWM controller 2 in response to voltage feedback V_(FB)from the converter's output present across filter capacitor 8.Break-before-make, i.e. BBM, operation is needed to prevent shorting outoutput capacitor 8.

The synchronous rectifier MOSFET 7, which may be N-channel or P-channel,is considered floating in the sense that its source and drain terminalsare not permanently connected to any supply rail, i.e. neither to groundor V_(batt). Diode 6 is a P-N diode intrinsic to synchronous rectifierMOSFET 7, regardless whether synchronous rectifier is a P-channel or anN-channel device. A Schottky diode may be included in parallel withMOSFET 7 but with series inductance may not operate fast enough todivert current from forward biasing intrinsic diode 6. Diode 9 comprisesa P-N junction diode intrinsic to N-channel low-side MOSFET 4 andremains reverse biased under normal boost converter operation. Sincediode 7 does not conduct under normal boost operation, it is shown asdotted lines.

If we define the converter's duty factor D as the time that energy flowsfrom the battery or power source into the DC/DC converter, i.e. duringthe time that low-side MOSFET switch 4 is on and inductor 3 is beingmagnetized, then the output to input voltage ratio of a boost converteris proportionate to the inverse of 1 minus its duty factor, i.e.

$\frac{V_{out}}{V_{in}} = {\frac{1}{1 - D} \equiv \frac{1}{1 - {t_{sw}/T}}}$

While this equation describes a wide range of conversion ratios, theboost converter cannot smoothly approach a unity transfer characteristicwithout requiring extremely fast devices and circuit response times. Forhigh duty factors and conversion ratios, the inductor conducts largespikes of current and degrades efficiency. Considering these factors,boost converter duty factors are practically limited to the range of 5%to 75%.

The Need for Dual Polarity Regulated Voltages: Today's electronicdevices require a large number of regulated voltages to operate, some ofwhich may be negative with respect to ground. Some smart phones may usemore than twenty-five separate regulated supplies in a single handheld,including negative bias supply needed for some organic light emittingdiodes (OLEDs), displays, for biasing LCD's, and for a variety of otherapplications. Space limitations preclude the use of so many switchingregulators each with separate inductors.

Unfortunately, multiple output non-isolated converters capable ofgenerating both positive and negative supply voltage require multiplewinding or tapped inductors. While smaller than isolated converters andtransformers, tapped inductors are also substantially larger and tallerin height than single winding inductors, and suffer from increasedparasitic effects and radiated noise. As a result multiple windinginductors are typically not employed in any space sensitive or portabledevice such as handsets and portable consumer electronics.

As a compromise, today's portable devices employ only a few switchingregulators in combination with a number of linear regulators to producethe requisite number of independent supply voltages. While theefficiency of the low-drop-out linear regulators, or LDOs, is oftenworse than the switching regulators, they are much smaller and lower incost since no coil is required. As a result efficiency and battery lifeis sacrificed for lower cost and smaller size. Negative supply voltagesrequire a dedicated switching regulator that cannot be shared withpositive voltage regulators. More than one negative regulated supplyvoltage may be required.

What is needed is a switching regulator implementation capable ofproducing both multiple positive and negative outputs, i.e. multipledual polarity outputs, from a single winding inductor, minimizing bothcost and size.

SUMMARY OF THE INVENTION

This disclosure describes an inventive boost converter able to producemultiple independently-regulated outputs of opposite polarity, i.e. oneor more positive above-ground output and one or more negativebelow-ground output from one single-winding inductor. A representativeimplementation of the dual polarity inductive boost converter includesan inductor and a switching network, the switching network configured toprovide the following modes of circuit operation: 1) a first mode wherethe positive electrode of the inductor is connected to an input voltageand the negative electrode of the inductor is connected to ground; 2) asecond mode where the negative electrode of the inductor in sequence toone or more of a first, second and third output nodes and the positiveelectrode of the inductor is connected in sequence to one or more of afourth and fifth output nodes; and 3) a third mode where the positiveelectrode of the inductor is connected to the input voltage and thenegative electrode of the inductor is connected to the either the third,fourth, or fifth output node output node. For clarification, it shouldbe noted that said positive electrode so named because it has a higherpositive potential during charging in the first mode of operationactually exhibits a negative potential during the second mode ofoperation. Said negative terminal of the inductor, while having apotential during magnetizing more negative than the positive terminal ofthe inductor, during the second and third modes of operation exhibits amore positive voltage than the inductor's other terminal.

The first mode of operation charges, i.e. magnetizes, the inductor to avoltage roughly equal to the input voltage. The second mode of operationtransfers charge to the first or second output nodes whilesimultaneously transferring charge to the third, fourth, or fifth outputnodes.

During the second mode of operation, in one embodiment of the inventionafter charge is transferred to the first output node, charge transferfrom the inductor is diverted to the second output node while thecircuit remains in its second mode of operation. During charge transferof the second operating mode, first and second output nodes becomebiased to negative voltages i.e. below ground, potentials. In tandem tothe sequential charging of the first and second output nodes, charge isalso transferred to a third output node, followed sequentially by thecharging of a fourth and optionally by a fifth output node. Duringcharge transfer of the second operating mode, third, fourth andoptionally fifth output nodes become biased to positive boosted voltagesi.e. above the converter's input voltage.

Once the second or the fifth output node reaches its target voltage, theconverter's second operating mode ends. Assuming the second output nodereaches its target voltage the third mode of operation continuescharging the third, fourth and fifth output nodes in sequence until thefifth reaches its target voltage. In this way, the boost converterprovides five regulated outputs from a single inductor, the chargingtime of each output node used to determine the value of the output.

It will be obvious to anyone skilled in the art that this technique canbe scaled to a fewer or greater number of positive and negative outputchannels.

For a second embodiment, the same basic components are used. In thiscase, however, the switching network provides the following modes ofoperation: 1) a first mode where the positive electrode of the inductoris connected to an input voltage and the negative electrode of theinductor is connected to ground; 2) a second mode where the positiveelectrode of the inductor is connected to the input voltage and thenegative electrode of the inductor is connected to either a third,fourth or fifth output node; and 3) a third mode where the positiveelectrode of the inductor is connected to the first or second outputnode and the negative electrode of the inductor is connected to ground.

The first mode of operation charges the inductor to a voltage equal tothe input voltage. The second mode of operation transfers chargesequentially to the third, fourth and fifth output nodes and ends whenthe fifth output node reaches a target voltage. The third mode ofoperation transfers charge sequentially to the first and second outputnodes; and ends when second output node reaches its target voltage. Inthis way, the boost converter provides five regulated outputs from asingle inductor.

In a third embodiment, the converter alternates between operation inaccordance with the first and second embodiments depending on whichoutput voltage requires a longer duration to reach or maintain atargeted output voltage.

In the first three embodiments, both positive and negative outputs arecharged in tandem until one of the two outputs reaches the third state.In an alternative implementation, the two outputs are charged notsimultaneously but in alternating sequence.

In an alternative embodiment the sequence and on time of charging thepositive and negative outputs varies algorithmically in response to theoutput voltages staying within a predetermined tolerance range of thetargeted output voltages.

In another embodiment the sequencing of the power MOSFETs is controlledalgorithmically by a microprocessor or other programmable logic inresponse to feedback from one or more analog-to-digital convertersmonitoring the output voltages.

In one embodiment the power MOSFETs connecting the inductor to thepositive outputs utilize either P-channel or N-channel MOSFETs withcircuitry to prevent forward biasing of any diode between the MOSFETssource and drain terminals. In a preferred embodiment the MOSFETconnected to the highest positive output voltage includes a source-bodyshort and an intrinsic diode parallel to its source and drain terminals.

In another embodiment, the power MOSFETs connecting the inductor to thepositive outputs utilize P-channel MOSFETs with the body connection tiedto a fixed positive potential, preferably the most positive outputvoltage.

In one embodiment the power MOSFETs connecting the inductor to thenegative outputs utilize either P-channel or N-channel MOSFETs withcircuitry to prevent forward biasing of any diode between the MOSFETssource and drain terminals. In a preferred embodiment the MOSFETconnected to the most negative output voltage includes a source-bodyshort and an intrinsic diode parallel to its source and drain terminals.

In another embodiment, the power MOSFETs connecting the inductor to thenegative outputs utilize isolated N-channel MOSFETs with the bodyconnection tied to a fixed positive potential, preferably the mostnegative output voltage and with an isolation region connected to themost positive output voltage or alternatively, the converter's input

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of a prior art single output synchronous boostconverter.

FIG. 2 is a schematic of a dual-polarity five-output synchronous boostconverter as provided by the present invention.

FIGS. 3A-3D show the boost converter of FIG. 2 performing an operationalsequence that implements a mode referred to as synchronous transfer.Synchronous transfer mode includes the following example successiveoperational phases: the inductor is magnetized (3A), charge issynchronously transferred to both +V_(OUT1) and to −V_(OUT4) (3B),charge is synchronously transferred to both +V_(OUT2) and to −V_(OUT5)(3C), and finally charge continues to be transferred exclusively to+V_(OUT3) (3D).

FIG. 4 comprises current and voltage plot of switching-waveformscharacteristic of the boost converter of FIG. 2 operating in synchronoustransfer mode.

FIG. 5 is a flowchart for the boost converter of FIG. 2 usingsynchronous transfer mode.

FIG. 6 illustrates the switching waveform of the boost converter of FIG.2 showing time multiplexed charging on the positive output nodes andtheir related equivalent circuits.

FIGS. 7 illustrates a another embodiment of the switching waveform ofthe boost converter of FIG. 2 showing an alternative time multiplexedsequence charging on the positive output nodes.

FIG. 8 illustrates one implementation of the boost converter of FIG. 2using P-channel MOSFETs as synchronous rectifiers with adaptive bodybias circuitry to supply the positive output nodes.

FIG. 9 illustrates an alternate implementation of the boost converter ofFIG. 2 using P-channel MOSFETs as synchronous rectifiers to supply thepositive output nodes without requiring adaptive body bias circuitry.

FIG. 10 illustrates an integrated circuit cross section showing oneembodiment for monolithic integrating P-channel synchronous rectifiers.

FIG. 11 illustrates one implementation of the boost converter of FIG. 2using N-channel MOSFETs as synchronous rectifiers with adaptive bodybias circuitry to supply the negative output nodes.

FIG. 12 illustrates an alternate implementation of the boost converterof FIG. 2 using N-channel MOSFETs as synchronous rectifiers to supplythe negative outputs nodes without requiring adaptive body biascircuitry.

FIG. 13 illustrates an integrated circuit cross section showing oneembodiment for monolithic integrating N-channel synchronous rectifiers.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

As described previously, conventional non-isolated switching regulatorsrequire one single-winding inductor and corresponding dedicated PWMcontroller for each regulated output voltage and polarity. In contrast,this disclosure describes an inventive boost converter able to producemultiple independently-regulated outputs of opposite polarity, i.e. oneor more positive above-ground outputs and one or more negativebelow-ground output from one single-winding inductor.

Shown in FIG. 2, a five-output dual polarity inductive boost converter10 comprises low-side N-channel MOSFET 11, inductor 12, high-sideP-channel MOSFET 13, floating positive-output synchronous rectifier 23with intrinsic source-to-drain diode 26, floating positive-outputsynchronous rectifiers 22 and 21 with no parallel source-drain diodes,floating negative-output synchronous rectifier 25 with intrinsicsource-to-drain diode 27, floating negative-output synchronous rectifier24 with no parallel source-drain diode, output filter capacitors 31, 32,33, 34 and 35 filtering outputs +V_(OUT1), +V_(OUT2), +V_(OUT3),−V_(OUT4), and −V_(OUT5). Regulator operation is controlled byPWM-controller 16 including break-before-make gate buffers 17 and 18,which control the on-time of MOSFETs 11, 13, 21, 22, 23, 24 and 25. PWMcontroller 16 may operate at fixed or variable frequency.

Closed-loop regulation is achieved through independent feedback from theoutputs +V_(OUT1), +V_(OUT2), +V_(OUT3), −V_(OUT4), and −V_(OUT5) usingcorresponding feedback signals V_(FB1), V_(FB2), V_(FB3), V_(FB4), andV_(FB5). The feedback voltages may be scaled by resistor dividers (notshown) or other level shift circuitry as needed. Low-side MOSFET 11includes intrinsic P-N diode 15 shown by dotted lines, which undernormal operation remains reverse biased and non-conducting. Similarly,high-side MOSFET 13 includes intrinsic P-N diode 14 shown by dottedlines, which under normal operation remains reverse biased andnon-conducting. High-side MOSFET 13 may be implemented using eitherP-channel or N-channel MOSFETs with appropriate adjustments in gatedrive circuitry.

Unlike in conventional boost converters, in dual-polarity boostconverter 10 magnetizing the inductor requires turning on both ahigh-side MOSFET 13 and a low-side MOSFET 11. Inductor 12 is thereforenot hard-wired to either V_(batt) or to ground. As a result theinductor's terminal voltages at nodes V_(x) and V_(y) are notpermanently fixed or limited to any given voltage potential except byforward biasing of intrinsic P-N diodes 14 and 15 and by the avalanchebreakdown voltages of the devices employed.

Specifically, node V_(y) cannot exceed one forward-biased diode dropV_(f) above the battery input V_(batt) without forward biasing P-N diode14 and being clamped to a voltage (V_(batt)+V_(f)). In the disclosedconverter 10, inductor 12 cannot drive the V_(y) node voltage aboveV_(batt), so that only switching noise can cause diode 14 to becomeforward biased.

Within the specified operating voltage range of the related devices,however, V_(y) can operate at voltages less positive than V_(batt) andcan even operate at voltages below ground, i.e. V_(y) can operate atnegative potentials.

The most negative V_(y) potential is limited by the BV_(DSS7) breakdownof high-side MOSFET 13, a voltage corresponding to the reverse biasavalanche of intrinsic P-N diode 14. To avoid breakdown, the MOSFET'sbreakdown must exceed the maximum difference between V_(y), which may benegative, and V_(batt), i.e. BV_(DSS7)>(V_(batt)−V_(y)). The maximumoperating voltage range of V_(y) is then bounded by the breakdown andforward biasing of diode 14 given by the relation

(V _(batt) +V _(f))>V _(y)>(V _(batt) −BV _(DSS7))

Similarly, node V_(x) cannot be biased beyond one forward-biased diodedrop V_(f) below ground without forward biasing P-N diode 15 and beingclamped to a voltage V_(x)=−V_(f). In the disclosed converter 10,however, inductor 12 cannot drive the V_(x) node voltage below ground,so that only switching noise can cause diode 15 to become forwardbiased.

Within the specified operating voltage range of the related devices,however, V_(x) can operate at voltages above ground and typicallyoperates at voltages more positive than V_(batt). The most positiveV_(x) potential is limited by the BV_(DSS6) breakdown of low-side MOSFET11, a voltage corresponding to the reverse bias avalanche of intrinsicP-N diode 15. To avoid breakdown, the MOSFET's BV_(DSS7) breakdown mustthe maximum of positive voltage of V_(x), which should exceed V_(batt),i.e. BV_(DSS6)>V_(x). The maximum operating voltage range of V_(x) isthen bounded by the breakdown and forward biasing of diode 15 given bythe relation

BV _(DSS6) >V _(x)>(−V _(f))

With the V_(y) terminal of inductor 12 being able to operate at voltagebelow ground and the V_(x) terminal of inductor 12 being able to operateabove V_(batt), the circuit topology of disclosed dual-polarity boostconverter 10 is significantly different than conventional boostconverter 1 which can only operate above ground and has its inductorhard wired to its positive input voltage. Since inductor 12 is nothard-wired to any supply rail, the disclosed dual-polarity boostconverter can therefore be considered a “floating inductor” switchingconverter. A conventional boost converter is not a floating inductortopology.

Operation of the disclosed dual-polarity boost converter involvesalternating between magnetizing the inductor and then transferringenergy to the outputs, before magnetizing the inductor again. Energyfrom the inductor may be transferred to both positive and negativeoutputs simultaneously or in alternating fashion.

Whenever energy is transferred to the positive outputs, the inductorcurrent is multiplexed sequentially to the positive outputs +V_(OUT1),+V_(OUT2), and +V_(OUT3) charging each output completely or partially totheir target voltages. The sequencing of the positive outputs may occurin any order independent of the charging of the converter's negativeoutputs.

Similarly, whenever energy is transferred to the converter's negativeoutputs, the inductor current is multiplexed sequentially to thenegative Outputs −V_(OUT4) and −V_(OUT5) charging each output completelyor partially to their target voltages. The sequencing of the negativeoutputs may occur in any order independent of the charging of theconverter's positive outputs.

Regardless of the algorithm employed for time multiplexing the variousoutputs, the first step in the operation of the disclosed dual-polarityboost converter is to store energy in, or herein to “magnetize”, theinductor, a process similar to charging a capacitor except the energy isstored in a magnetic rather an electric field.

Inductor Magnetizing, FIG. 3A illustrates operation 40 of converter 10during the magnetizing of inductor 12. Since inductor 12 is connected tobattery input V_(batt) through not one, but two series connectedMOSFETs, then both low-side and high-side MOSFETs 11 and 13 must beturned on simultaneously to allow current I_(L)(t) to ramp. Meanwhilesynchronous rectifier MOSFETs 21 through 25 remain off andnon-conducting. The current-voltage relationship for an inductor isgiven by the differential equation

$V_{L} = {L\frac{I}{t}}$

which for small intervals can be approximated by the difference equation

$V_{L} \cong {L\; \frac{\Delta \; I}{\Delta \; t}}$

Assuming minimal voltage drop across on-state MOSFETs 11 and 13, thenV_(L)≈V_(batt) and the above equation can be rearranged as

$\frac{\Delta \; I}{\Delta \; t} = {\frac{V_{L}}{L} \approx \frac{V_{batt}}{L}}$

which describes for short magnetizing intervals the current I_(L)(t) ininductor 12 can be approximated as a linear ramp of current with time.For example as shown in graph 90 of FIG. 4, during the interval betweento and t₁ the current I_(L) ramps linearly 91 from some non-zero currentat time t₀ toward a peak value 92 at time t₁, the end of the magnetizingoperating phase. The energy stored in inductor 12 at any time t is givenby

${E_{L}(t)} = \frac{{LI}^{2}(t)}{2}$

reaching its peak E_(L)(t₁) just before its current is interrupted byswitching off one or both MOSFETs 11 and 13. As shown in graphs 60, 75and 90 of FIG. 4, during magnetizing the current 16 in low-side MOSFET11 and the current 17 in high-side MOSFET 13 are identical and equal tothe inductor current I_(L) so that in the interval t₀ to t₁,

I ₆(t)=I ₇(t)=I _(L)(t)

At current I₆(t), a small voltage drop V_(DS2(on)) appears acrossseries-connected low-side N-channel MOSFET 11. Operating in its linearregion and carrying current I_(L)(t) with an on-state resistance ofR_(DS6(on)) the voltage V_(x) is given by

V _(x) =V _(DS6(on)) =I _(L) ·R _(DS6(on))

as shown by line 61 in graph 60 of FIG. 4. For low on-resistances,typically a few hundred milliohms or less, then V_(x) is approximatelyequal to ground potential, i.e. V_(x)≈0. Similarly, a small voltage dropV_(DS7(on)) also appears across series-connected high-side P-channelMOSFET 13. Operating in its linear region at a current I_(L)(t) with anon-state resistance of R_(DS7(on)) the voltage V_(y) is then given by

V _(y) =V _(batt) −V _(DS7(on)) =V _(batt) −I _(L) ·R _(DS7(on))

as shown by line 52 in graph 75 of FIG. 4. For low on-resistances, thenV_(y) is approximately equal to the battery potential, i.e.V_(y)≈V_(batt).

Given that V_(x)≈0 and V_(y)≈V_(batt) then the approximationV_(L)=(V_(y)−V_(x))≈V_(batt) is a valid assumption. Accordingly, theramp in inductor current shown in graph 90 can, as described previously,therefore be approximated as a straight line segment with a slope(V_(batt)/L). Furthermore assuming the voltage +V_(OUT3) acrosscapacitor 31 is above ground and the voltage −V_(OUT5) across capacitor35 is below ground, then +V_(OUT3)>V_(x) and V_(y)>−V_(OUT5) so that P-Ndiodes 26 and 27 are both reverse biased and non-conducting.

Synchronous Energy Transfer to Dual Polarity Outputs: After magnetizinginductor 12, in the synchronous transfer algorithm both low-side andhigh-side MOSFETs 11 and 13 are turned off simultaneously, as shown attime t₁ of FIG. 4. Interrupting the I₇ current in high-side MOSFET 13and the I₆ current in low-side MOSFET 11 causes the inductor's V_(x)terminal to fly up to a positive voltage 63 greater than V_(OUT1),forward biasing diode 26. Before any appreciable energy flows tocapacitor 33 and output +V_(OUT3), MOSFET 21 turns on and reroutes thecurrent to capacitor 31 and output +V_(OUT1) as shown in FIG. 3B. Thevoltage V_(X) then settles to a voltage 63 only slightly above+V_(OUT1), the product of the inductor current and the on-resistance ofsynchronous rectifier MOSFET21. i.e. I_(L)R_(DS1).

As shown in FIG. 3B synchronous to inductor 12 transferring energy tocapacitor 31, the same inductor also charges capacitor 34 to a negativepotential. Specifically at time t=t₁ as shown in graph 75 of FIG. 4 theinductor's V_(y) terminal voltage flies down to a below-ground voltage80, a voltage more negative than −V_(OUT5), momentarily forward biasingdiode 27. Before any appreciable energy is transferred to capacitor 35,MOSFET 24 is turned on reroutes the inductor's current to the negativevoltage output −V_(OUT4). The voltage V_(Y) then settles to a voltage 81only slightly below −V_(OUT4), the product of the inductor current andthe on-resistance of synchronous rectifier MOSFET 24. i.e. I_(L)R_(DS4).

During the transition at time t₁, break-before-make circuits 17 and 18prevents synchronous rectifier MOSFETs 21 and 24 from turning on andmomentarily shorting out filter capacitors 31 and 34. Without MOSFETconduction, diodes 26 and 27 carry the inductor current I_(L) andexhibit a forward-biased voltage-drop V_(f). In the absence of straycapacitance in the circuit, the instantaneous voltage 62 on V_(x) duringthe BBM interval is then equal to (V_(OUT3)+V_(f)). The instantaneousvoltage 80 on V_(y) is similarly equal to (−V_(OUT5)−V_(f)). If however,the BBM interval is sufficiently short, stray capacitance filters thevoltage spikes 62 and 80 on V_(X) and V_(Y) to a magnitude slightlylarger than +V_(OUT1) and −V_(OUT4). As a result, no significant energyflows to capacitors 33 and 35 during the BBM interval and the outputvoltages +V_(OUT3) and −V_(OUT5) are not disturbed.

Since no significant power flows through them, the function of diodes 26and 27 is to act as a clamp to prevent short unwanted spikes and relatednoise on the V_(X) and V_(Y) and to avoid driving MOSFETs 11 and 13 intoa potentially harmful unclamped-inductively-switched avalanchecondition. In graphs 60 and 75 theses short BBM intervals arerepresented as temporary V_(X) voltage transients 62, 68, 69, 70 andtemporary V_(Y) voltage transients 80, 84 and 85.

After the BBM interval at time t₁, the positive and negative outputs+V_(OUT1) and −V_(OUT4) charge simultaneously during which inductor 12essentially decouples the voltage at nodes V_(x) and V_(y) allowing themto act independently during the time energy is transferred to the loadsand to output capacitors 31 and 34 through the synchronous rectifierMOSFETs 21 and 24.

In one method, the condition shown in schematic 3B should continue untilthe voltage on either capacitor 31 or 34 comes into a specifiedtolerance range. The tolerance range of the target voltage is determinedby the controller in response to the feedback signals V_(FB1) andV_(FB4). Using analog control, the PWM controller 16 includes an erroramplifier, a ramp generator, and a comparator to determine when to shutoff each synchronous rectifier. Using digital control, this decision canbe made by logic or software according a specified algorithm, dependingon the load current demand and capacitor voltage on each output.

For example when the positive output +V_(OUT1) reaches its targetvoltage at time t₂, positive synchronous rectifier MOSFET 21 is turnedoff discontinuing charging of capacitor 31. After a BBM interval MOSFET22 is turned on and capacitor 32 then commences charging of outputvoltage +V_(OUT2). The voltage V_(X) then increases to(+V_(OUT2)+I_(L)R_(DS2)) as shown by line 65 in graph 60.

At a later time, i.e. at time t₃, the negative output −V_(OUT4) reachesits specified tolerance range, synchronous rectifier MOSFET 24 is turnedoff. After a BBM interval, MOSFET 25 is turned on and capacitor 35commences charging of output −V_(OUT5). Charging of capacitor 32 andvoltage +V_(OUT2) continues. This condition 50 from time t₄ to t₅ isillustrated in FIG. 3C during which the voltage V_(X) then increases to(+V_(OUT3)+I_(L)R_(DS3)) as shown by line 75 in graph 60 and where V_(Y)increases to (−V_(OUT5)−I_(L)R_(DS5)) as shown by line 83 in graph 75.

Synchronous Energy Transfer to One Polarity Output: Depending on loadconditions either positive or negative polarity outputs may become fullycharged within their tolerance ranges first. Once either output reachesits specified output voltage, the converter is again reconfigured todiscontinue charging of the fully charged polarity but continue chargingthe output capacitors not yet within the tolerance range its specifiedvoltage target.

For example, if at a time t₅ the negative output −V_(OUT5) reaches itstarget voltage before +V_(OUT3), then the first action is to turn offsynchronous rectifier MOSFET 25, and disconnect capacitor 35 from overcharging. After BBM interval 59 is completed, high-side MOSFET 13 isturned-on and V_(y) jumps to a voltage of V_(batt)−I_(L)·R_(DS7(on))shown by line 56 in graph 60. During the hand-off at time t₅, inductorcurrent I_(L) is diverted from I₅ to I₇ in the transition shown by point84 in graph 74. Current I₃ however remains unchanged.

This condition is shown in circuit 55 of FIG. 3D where the current pathof I_(L) flows from V_(batt) through conducting high-side MOSFET 13,inductor 12, and on-state positive synchronous rectifier 23 so thatI_(L)=I₇=I₃. Capacitor 33 therefore continues to charge even thoughcharging of capacitor 35 has stopped. With V_(y) biased near V_(batt)and −V_(OUT5) below ground P-N diode 27 remains reversed biased andnon-conducting.

The operating phase of circuit 35 is maintained until +V_(OUT3) reachesits target voltage at time T. Once +V_(OUT3) is at its target voltage,positive synchronous rectifier MOSFET 23 is turned off and for thebreak-before-make duration t_(BBM) 68, diode 26 carries the inductorcurrent.

Once however the BBM interval 68 is completed low-side MOSFET 11 isturned on, current is diverted from I₃ to I₆ and inductor 12 begins anew cycle of being magnetized returning to the state shown in circuit40. Having completed the cycle, the total time is described as theperiod T which will vary depending on load current. This period isdetermined by the magnetizing duration and the positive or negativecharge transfer phases which ever is longer.

The example given in FIG. 3D described a case where the negative Output−V_(OUT5) reached its target voltage before the positive Output+V_(OUT3). The converter also accommodates the opposite scenario, i.e.when the positive voltage hits its point of regulation first.

As shown in graph 90 in the synchronous transfer method, the inductor 12must supply all five outputs with charge over an interval from t₁ to Twith the inductor current decaying 93 from a peak current 92 to aminimum value 94 before the cycle repeats. For conservation of energy,the magnetizing energy during the interval t₀ to t₁ must equal theenergy delivered in the remainder of the period.

State Diagram of Synchronous Charge Transfer The algorithm and statediagram for synchronous transfer 100 is illustrated in FIG. 5. As shownthe initial state 110 involves magnetizing the inductor, thensimultaneously operations 101 and 106 power both negative and positiveoutputs by turning off the high side and low side MOSFETs 11 and 13 andtime multiplexing the synchronous rectifiers. In flow 101, 102, 103 thepositive polarity outputs +V_(OUT51), +V_(OUT2), and +V_(OUT3) arecharged 111 sequentially as shown or in any sequence. In tandem tocharging the positive outputs, in flow 106, 107 the converter's negativeoutputs are charged in any sequence.

The charging of the converter's positive outputs is controlled by thelow-side MOSFET 11 connected between ground and V_(X). Turning offMOSFET 11 commences charging according to multiplexed sequence 101, 102,and 103. To terminate positive charging 104, low side MOSFET 11 must beturned back on in state 112. The result is conditional. If high sideMOSFET 13 is already on, then turning on low side MOSFET 112 willre-initiate 105 magnetizing inductor 12 shown by state 110. If the highside MOSFET is still off, i.e. if negative charging sequence 106, 107 isstill ongoing, then the positive loop will wait in condition 112.

Similarly, the charging of the converter's negative outputs iscontrolled by the high-side MOSFET 13 connected between V_(batt) andV_(Y). Turning off MOSFET 13 commences charging of the negative outputs113 according to multiplexed sequence 106 and 107. To terminate negativecharging 108, high side MOSFET 13 must be turned back on in state 114.The result is conditional. If low side MOSFET 11 is already on, thenturning on low side MOSFET 115 will re-initiate 109 magnetizing inductor12 shown by state 110. If the low side MOSFET is still off, i.e. if thepositive charging sequence 101, 102, 103 is still ongoing, then thenegative loop will wait in condition 114.

In the synchronous transfer method, both loops in algorithm 100 occursimultaneously, the longer loop sets the duration of the repeatedinterval, i.e. the converter's period T. For example if the negativecharge transfer sequence 106, 107, 108 occurs in a shorter time than thepositive loop 101, 102, 103, 104, the negative loop will wait at state114 with its high side MOSFET on until the positive loop reaches state112. When state 112 finally is reached, then the converter returns tothe starting condition by paths 105 and 109 simultaneously.

Conversely, if the positive charge transfer sequence 101, 102, 103, 104occurs in a shorter time than the negative loop 106, 107, 108, thepositive loop will wait at state 112 with its low side MOSFET on untilthe negative loop reaches state 114. When state 114 finally is reached,then the converter returns to the starting condition by paths 105 and109 simultaneously.

In the disclosed approach, charging is synchronous because both highside and low side MOSFETs are turned off simultaneously therebyimmediately forcing both sides of the inductor V_(X) and V_(Y) toexhibit voltage transients an charging their respective outputs.

An alternative approach is to alternate between the positive and thenegative loops, first by magnetizing the inductor, turning off only thelow side MOSFET and completing positive output loop 101, 102, 103, 104,105, returning to magnetizing state 110, turning off only the high sideMOSFET, completing negative output loop 106, 107, 108, 109, and thenrepeating the entire process.

Voltage Regulation of the Dual-Polarity Multiple Output Regulator:Operation of the dual polarity boost converter requires turning on bothhigh-side and low-side MOSFETs 13 and 11 to magnetize inductor 12 andthen shutting off these MOSFETs to transfer energy to the convertersoutputs. In the synchronous energy transfer algorithm, bothaforementioned high-side and low-side MOSFETs are shut offsimultaneously starting the transfer of energy from the inductor to bothoutputs simultaneously.

Despite being charged synchronously, independent regulation of thepositive and negative outputs are determined by the duration of energytransfer to each output. Specifically, by controlling the off-time ofthe low-side and high-side MOSFETs 11 and 14 and the relative on-timefor each of the synchronous rectifier MOSFETs through feedback, thevarious positive and negative output voltages may be independentlyregulated from a single inductor.

Time Multiplexed Sequencing In the disclosed invention, any timemultiplexed sequence may be used for producing the multiple positive ormultiple negative outputs. For example in FIG. 6, the positive outputsare charged in successive monotonic sequence starting with the lowestoutput voltage +V_(OUT1), progressing to the second output +V_(OUT2),and finally charging the highest output voltage +V_(OUT3). The graphillustrates the initial charging of the output capacitors during startup, not just steady state operation.

More specifically from time t₀ to t₁ sub-circuit 140 illustrates thelow-side MOSFET 11 is on and the synchronous rectifiers are off. Allthree output voltages 131 in graph 130 are at zero and V_(X) in graph120 has a potential 121 equal to the voltage drop I_(L)R_(DS(on)) acrossthe conducting low side MOSFET 11.

After MOSFET 11 is shut off at time t₁, synchronous rectifier 21 shownin sub-circuit 141 is turned on and +V_(OUT1) ramps 132 to its targetvoltage V′_(OUT1). At the same time diode 26 becomes forward biasedramping +V_(OUT3) to value of (+V_(OUT1)−V_(f)). The inductor nodevoltage V_(X) drives the outputs voltage up with a ramp 122 limited bythe charging of the converter's output capacitances. During this period,+V_(OUT2) remained at ground.

At time t₂ MOSFET 21 is turned off and MOSFET 22 is turned on as shownin sub-circuit 142. As V_(x) continues 123 to rise, the output voltage+V_(OUT2) charges to a target voltage V′_(OUT2) and the forward biasingof diode 26 continues to ramp output +V_(OUT3) to value of(+V_(OUT2)−V_(f)) as shown by line 135. With MOSFET 21 off, the output+V_(OUT1) remains constant at its targeted value V′_(OUT1).

At time t₃ MOSFET 22 is turned off and MOSFET 23 is turned on as shownin sub-circuit 143. As V_(x) continues 124 to rise, the output voltage+V_(OUT3) charges 138 to a target voltage V′_(OUT3) with MOSFET 23shunting forward biased diode 26. With MOSFETs 21 and 22 off, the lowervoltage outputs +V_(OUT1) and +V_(OUT2) remain constant at or near theirtargeted values V′_(OUT1) and V′_(OUT2).

After time T, the circuit enters steady state operation with only smallchanges in the converter's outputs manifest thereafter.

An alternative multiplexing sequence, one where the V_(X) voltage doesnot ramp monotonically, is illustrated in FIG. 7 where graph 150illustrates V_(X) and graph 160 illustrates the various output voltages.Specifically, before time t₁, the inductor is magnetizing, V_(X) isbiased near ground, and all three outputs are zero.

At time t₁, the low side MOSFET is turned off and MOSFET 23 is turnedon, whereby the converter's highest output-voltage +V_(OUT3) rises 162toward its target value V′_(OUT3) driving by V_(X) voltage 152. Otheroutputs +V_(OUT1) and +V_(OUT2) remain grounded.

At time t₂, the converter's lowest output voltage +V_(OUT1) is nextpowered ramping 163 in proportion to V_(X) 153 toward a target value ofV′_(OUT1). The output +V_(OUT2) remains grounded. Because the output+V_(OUT3) is fully charged, P-N diode 26 becomes reverse biased.

At time t₃ the converter's median output voltage +V_(OUT2) is nextpowered ramping 164 in proportion to V_(X) 154 toward a target value ofV′_(OUT2). The outputs +V_(OUT1) and +V_(OUT1) remain at their previousvalues. Because the output +V_(OUT3) is fully charged, P-N diode 26remains reverse biased.

So the disclosed converter's initial multiplexing sequence can beimplemented without concern for special sequencing for monotonicoperation.

In the example of circuit 10 operation, in steady state operation onlyone positive and one negative synchronous rectifier are turned on at atime. Specifically as shown whenever low-side MOSFET 11 is biased offand the voltage at node V_(X) increases, only one positive-outputconnected synchronous rectifier MOSFET, either MOSFET 21, 22 or 23 isturned on at the same time. Similarly, whenever high-side MOSFET 13 isturned off and V_(Y) flies negative, only one negative-output connectedsynchronous rectifier MOSFET, either MOSFET 24 or 25 is turned on at thesame time. Turning on more than one synchronous rectifier would ineffect short out the affected outputs and cause the voltages

For example simultaneously turning on synchronous rectifier MOSFETs 21,22 and 23 will cause the voltage to equilibrate among capacitors 31, 32and 33. The adverse effect of voltage equilibration is that charge flows“backwards” from the highest output voltage capacitor into the lowestvoltage capacitor lowering converter efficiency and increasing outputripple or causing noise spikes on the affected outputs. If, during suchas condition V_(OUT3)>V_(OUT2), then by simultaneously turning onMOSFETs 23 and 22, capacitor 32 would be charged by a combination ofboth inductor 12 current and capacitor 33 current causing V_(OUT2) torise and V_(OUT3). to drop in voltage. Energy redistribution among thefilter capacitors is less efficient than supplying new charge to theoutputs from inductor 12 current as need.

Similarly, simultaneously turning on synchronous rectifier MOSFETs 24and 25 will cause the voltage to equilibrate among capacitors 34 and 35.The adverse effect of voltage equilibration is that charge flows“backwards” from the highest, i.e. the most negative, output voltagecapacitor into the lowest voltage capacitor. Such currents lowerconverter efficiency and increase output ripple and noise spikes on theaffected outputs. If, during such as condition −V_(OUT5)<−V_(OUT4), thenby simultaneously turning on MOSFETs 24 and 25, capacitor 34 would becharged by a combination of both inductor 12 current and capacitor 35current causing −V_(OUT5) to rise and −V_(OUT3.) to drop, i.e. be comeless negative, in voltage. Energy redistribution among the filtercapacitors is less efficient than supplying new charge to the outputsfrom inductor 12 current as need.

Since the positive and negative outputs are connected to oppositeterminals V_(X) and V_(Y) of inductor 12, the selection or sequence ofpositive-output connected synchronous rectifiers places no limitationson which negative-output connected synchronous rectifier MOSFET isconducting in tandem, or vice versa. As long as only one positive-outputconnected MOSFET and only one negative output connected MOSFET areconducting, no intra-capacitor charge redistribution will occur andefficiency will not be lost.

Although turning on multiple synchronous rectifiers of the same polaritydoes not necessarily damage the devices of circuit 10 or preventregulation, it offers no technical merit and generally suffers a numberof the aforementioned problems.

One condition, however, allows the outputs to be shorted by multipleconducting synchronous rectifiers without causing charge redistributionlosses in the capacitors. This condition occurs during start up when thecapacitors are being charged for the first time. As long as thecapacitor voltages are similar all the synchronous rectifiers may beturned on in tandem and allow the inductor to simultaneously chargeevery positive and negative output. This process expedites turn on andramp up of the converter to its steady state. Once a given outputcapacitor reaches its target range, it is disconnected from the inductorwhile the other channels continue to charge. Once disconnected theisolated capacitor will quickly exhibit its own unique output voltageand thereafter may be not be reconnected in parallel with the otheroutputs without charge redistribution losses occurring.

Other Features of the Dual Polarity Multi-Output Converter One featureof the disclosed converter 10 is that since the inductor is floating,i.e. not permanently connected to a supply rail, turning on either thehigh-side or low-side MOSFETs 11 and 13 but not both can force thevoltage at V_(y) or V_(x) without magnetizing or increasing the currentin inductor 12. This is not possible for a conventional boost converterlike the one in FIG. 1 where a single MOSFET both controls the V_(x)voltage but also causes current conduction, magnetizing the inductor. Inother words in a conventional converter, controlling the inductorvoltage also causes additional and sometimes unwanted energy storage. Inthe disclosed converter, either V_(x) or V_(y) can be forced to a supplyvoltage without magnetizing the inductor.

Another consideration is the output voltage range of conventional boostconverter 1. If a P-N diode 6 is present across a synchronous rectifierMOSFET, the minimum output voltage for the boost converter's output isnecessarily V_(batt), because the diode forward biases pulling theoutput up to V_(batt) as soon as power is applied to the regulator'sinput terminals. In the disclosed dual output converter, the circuitpath from V_(X) to +V_(OUT1) or to +V_(OUT2) includes MOSFETs with nosource-drain parallel P-N diodes, allowing +V_(OUT1) or +V_(OUT2) toregulate a voltage less than V_(batt), a feature not possible with aconventional boost converter topology.

So while boost converters can only step up voltage, the disclosedconverter produces a positive output voltage that can be less than,equal to or greater than the battery voltage, and is therefore notrestricted to operation only above V_(batt). Adapting a boostconverter's topology for step-down voltage regulation is the subject ofa related patent by Richard K. Williams entitled “High-EfficiencyUp-Down and Related DC/DC Converters” (now U.S. patent application Ser.No. 11/835,809) and is included herein by reference.

In a related disclosure entitled “Dual-Polarity Multi-Output DC/DCConverters and Voltage Regulators” (now U.S. patent application Ser. No.11/890,818) by Richard K. Williams, the application of atime-multiplexed-inductor in both positive and negative output boostconverters is described and is incorporated herein by reference.

Multiplexer Implementation The disclosed dual polarity multi-outputconverter requires the use of MOSFETs free from parasiticsource-to-drain diodes. In order to implement power MOSFETs withoutintrinsic source to drain diodes, a number of methods are hereindisclosed. Once such method illustrated in sub-circuit 180 of FIG. 8comprises P-channel MOSFETs with integral body bias generator circuitry.

As shown P-channel MOSFETs 21 and 22 include associated body biasgenerator or BBG circuits 191A and 181A respectively

BBG circuit 191A comprises cross coupled MOSFETs 192 and 193 sharing acommon body connection to main P-channel MOSFET 21, the body whichrepresents the cathode of intrinsic diodes 194 and 195. In itsintegrated version BBG circuit 191A contained in an N-type well or tubmay include a parasitic diode 191B to ground. BBG circuit 181A is ofsimilar construction to 191A.

Operation of BBG circuit selectively shunts diodes 194 and 195 so thatneither diode can become forward biased and carry current regardless ofthe polarity of the source-drain terminals of MOSFET 21. For example ifV_(OUT1)>V_(X), then the gate of P-channel MOSFET 193 is more negativethan its other terminals so that P-channel 193 is on, shunting P-N diode195. The body of P-channel MOSFET 21 is therefore connected to V_(OUT1),the most positive device potential. Consequently, diode 194 remainsreversed biased and non-conducting. P-channel 192, with its gate to themost positive potential also remains off. Since the device and circuitare symmetric the argument applies equally in both polarities. As aresult BBG circuit 191A makes MOSFET 21 appear as if it has no parallelP-N diode that can ever become forward biased.

In circuit 180, while MOSFETs 21 and 22 use BBG circuits 191A and 181Ato prevent body diode conduction, MOSFET 23 connected to the mostpositive output voltage +V_(OUT3), does not require a BBG circuit. Infact parallel diode 26 is important as insurance to prevent V_(x) fromlarge voltage spikes.

In integrated form the N-well or epitaxial layers 181A and 191A formingthe body of the P-channel MOSFETs also form parasitic diodes 181B and191B to the surrounding P-type substrate. Since V_(X) is alwayspositive, these diodes remain reverse biased during normal operation ofbody bias generator circuitry 180.

A similar approach to circuit 180 can be employed using N-channel powerMOSFETs in place of P-channel devices but the gate drive circuitry mustbe modified accordingly for bootstrap circuitry with a floatingbootstrap capacitor. Generally a special wafer fabrication process isrequired to isolate N-channel MOSFETs from a surrounding P-typesubstrate. A P-type substrate is common in most wafer fabricationprocesses, especially in conventional CMOS processes.

Alternate P-channel solution 200 shown in FIG. 9 utilizes the knowledgethat the output +V_(OUT3) is the most positive potential and can be usedto avoid diode conduction in MOSFETs 21 and 22. As such P-channel MOSFET21 includes intrinsic body diodes 194 and 195 with cathode and N-typewell region 191A electrically tied to +V_(OUT3), the circuit's mostpositive potential. Similarly P-channel MOSFET 22 includes intrinsicbody diodes 184 and 185 with cathode and N-type well region 181Aelectrically tied to +V_(OUT3), the circuit's most positive potential.Because N-type wells 181A and 191A are biased at the highest potential,i.e. at +V_(OUT3), then in an integrated form parasitic diodes 181B and191B to the surrounding P-type substrate remain off and reversed-biasedduring normal circuit operation. One disadvantage of circuit 200compared to the BBG method of circuit 180 is the reverse biased sourceto body bias will result in an increase in threshold voltage andon-resistance.

Regardless of the circuit, the P-channel devices shown in schematics 180and 200 can be integrated using standard CMOS or preferably realized ina process that provides a heavily doped high, i.e. high concentration,buried layers. One such implementation illustrated in cross section 220of FIG. 10 utilizes a deep implanted N-type region 222 overlapping thebottom of N-type well 223. In the implementation shown well 223 and deepimplant 222 are formed conformal to LOCOS oxide 233. The P-channelMOSFET includes P+ source drain regions 224, P-type lightly doped drainextension 225, sidewall spacer 229, polysilicon gate 228, gate silicide227, contact barrier metal 235, 1^(st) layer metal 231 and second layermetal 232. This concept, referred to herein as a variable gate widthswitching converter, is described in prior art U.S. Pat. No. 5,973,367by Richard K. Williams and in another implementation in U.S. Pat. No.7,026,795 by John So.

As an example, in cross section 220 the N-type well shared by MOSFETs21, 22 and 23 are biased to the positive output potential +V_(OUT3).P-channel 21, not shown in cross section 229 can also be integrated intothe same shared well. If a BBG circuit such as shown in FIG. 8 isrequired, however MOSFETs 21, 22 and 23 must each employ their ownseparate wells and cannot share a common one.

The same BBG circuit technique can be applied for N-channel MOSFETs usedas synchronous rectifiers for the converter's negative outputs. As shownin circuit 250 of FIG. 11, MOSFET 24 includes cross coupled N-channelMOSFETs 254 and 255 to selectively shunt intrinsic diodes 253 and 252.Since the body of N-channel 24 is P-type it must be isolated from theP-type substrate in order to monolithically integrate circuit 250.Isolation of N-channel MOSFETs generally requires a special waferfabrication process whereby a N-type isolation layer 257A surroundingand enclosing the MOSFET forms a reversed biased diode 257B with thesurrounding P-type substrate and a reversed biased diode 257C with saidenclosed P-type well. N-type isolation layer 257A is biased at voltagemore positive than its operating voltages, e.g. to +V_(batt) or+V_(OUT3) to prevent forward biasing of the isolation region to itssurroundings. Floating N-channel MOSFET 25 also requires isolation sinceit is not ground connected, even though it contains source to draindiode 27 and a source-body short.

An alternative approach shown in circuit 270 of FIG. 12, biases theP-type body of N-channel MOSFET 24 to the most negative circuitpotential −V_(OUT5). The floating device still requires isolation fromthe P-type substrate using a specialized process with isolated devices.Formed in a P-well surrounded by an N-type isolation layer 271A, theisolation is generally biased to a potential V_(ISO) more positive thanthe operating voltage range of the device. Positive bias supplies maycomprise the V_(batt) input or a positive output such as +V_(OUT3).Properly biased, N-type isolation layer 271A forms a reversed biaseddiode 271B with its surrounding P-type material and also forms a reversebiased diode 271C with the P-well it encloses.

One such implementation is shown in cross section 300 with a deepimplanted N-isolation region 302 surrounding a P-type well 303containing two N-channel MOSFETs for producing outputs −V_(OUT4) and−V_(OUT5). The isolation is independently biased to a potential equal toor more positive that V_(batt). The N-channel MOSFET includes N+ sourcedrain regions 305, N-type lightly doped drain extension 306, sidewallspacer, gate oxide 307, polysilicon gate 309, gate silicide 310, contactbarrier metal 313, 1^(st) layer metal 312 and second layer metal (notshown). The process to fabricate such a device is also described in U.S.Pat. No. 6,855,985 “Modular Bipolar-CMOS-DMOS Analog Integrated Circuit& Power Transistor Technology” by Richard K. Williams et al.incorporated herein by reference.

1. A dual-polarity multi-output synchronous boost converter thatcomprises: an inductor; first, second, third, fourth and fifth outputnodes; and a switching network, the switching network configured toprovide the following modes of circuit operation: a first mode where thepositive electrode of the inductor is connected to an input voltage andthe negative electrode of the inductor is connected to ground; a secondmode where the negative electrode of the inductor is connected insequence to one or more of the first, second and third output nodes andthe positive electrode of the inductor is connected in sequence to oneor more of the fourth and fifth output nodes; and a third mode where thepositive electrode of the inductor is connected to the input voltage andthe negative electrode of the inductor is connected in sequence to oneor more of the first, second and third output nodes.
 2. A dual-polaritydual-output synchronous boost converter as recited in claim 2 thatfurther comprises a control circuit that causes the first, second andthird modes to be selected in a repeating sequence.
 3. A dual-polaritydual-output synchronous boost converter as recited in claim 3 in whichthe repeating sequence has the form: first mode, second mode, firstmode, third mode.
 4. A dual-polarity dual-output synchronous boostconverter as recited in claim 3 in which the repeating sequence has theform: first mode, second mode, third mode.
 5. A dual-polaritydual-output synchronous boost converter as recited in claim 1 thatfurther comprises a feedback circuit that modulates the duration of therespective connections between the inductor, the input voltage, groundand the output nodes to control the voltage of the output nodes.
 6. Adual-polarity dual-output synchronous boost converter as recited inclaim 1 in which the switching network is configured to provide a fourthmode where the negative electrode of the inductor is connected insequence to one or more of the first, second and third output nodes andthe positive electrode of the inductor is connected in sequence to oneor more of the fourth and fifth output nodes.
 7. A dual-polaritymulti-output synchronous boost converter that comprises: an inductor;first, second, third, fourth and fifth output nodes; and a switchingnetwork, the switching network configured to provide the following modesof circuit operation: a first mode where the positive electrode of theinductor is connected to an input voltage and the negative electrode ofthe inductor is connected to ground; a second mode the negativeelectrode of the inductor is connected to ground and the positiveelectrode of the inductor is connected in sequence to one or more of thefourth and fifth output nodes; and a third mode where the positiveelectrode of the inductor is connected to the input voltage and thenegative electrode of the inductor is connected in sequence to one ormore of the first, second and third output nodes.
 8. A dual-polaritydual-output synchronous boost converter as recited in claim 7 thatfurther comprises a control circuit that causes the first, second andthird modes to be selected in a repeating sequence.
 9. A dual-polaritydual-output synchronous boost converter as recited in claim 8 in whichthe repeating sequence has the form: first mode, second mode, firstmode, third mode.
 10. A dual-polarity dual-output synchronous boostconverter as recited in claim 8 in which the repeating sequence has theform: first mode, second mode, third mode.
 11. A dual-polaritydual-output synchronous boost converter as recited in claim 7 thatfurther comprises a feedback circuit that modulates the duration of therespective connections between the inductor, the input voltage, groundand the output nodes to control the voltage of the output nodes.
 12. Adual-polarity dual-output synchronous boost converter as recited inclaim 7 in which the switching network is configured to provide a fourthmode where the negative electrode of the inductor is connected to groundand the positive electrode of the inductor is connected in sequence toone or more of the fourth and fifth output nodes.
 13. A method foroperating a dual-polarity multi-output synchronous boost converter whichincludes an inductor and first, second, third, fourth and fifth outputnodes, the method comprising: configuring a switching network so thatthe boost converter operates in a first mode where the positiveelectrode of the inductor is connected to an input voltage and thenegative electrode of the inductor is connected to ground; configuringthe switching network so that the boost converter operates in a secondmode where the negative electrode of the inductor is connected to groundand the positive electrode of the inductor is connected in sequence toone or more of the fourth and fifth output nodes; and configuring theswitching network so that the boost converter operates in a third modewhere the positive electrode of the inductor is connected to the inputvoltage and the negative electrode of the inductor is connected insequence to one or more of the first, second and third output nodes. 14.A method as recited in claim 13 in which the first, second and thirdmodes are selected in a repeating sequence.
 15. A method as recited inclaim 14 in which the repeating sequence has the form: first mode,second mode, first mode, third mode.
 16. A method as recited in claim 14in which the repeating sequence has the form: first mode, second mode,third mode.
 17. A method as recited in claim 18 that further comprisesmodulating the duration of the respective connections between theinductor, the input voltage, ground and the output nodes to control thevoltage of the output nodes.
 18. A method as recited in claim 13 thatfurther comprises configuring the switching network so that the boostconverter operates in a fourth mode where the negative electrode of theinductor is connected to ground and the positive electrode of theinductor is connected in sequence to one or more of the fourth and fifthoutput nodes.
 19. A method for operating a dual-polarity multi-outputsynchronous boost converter which includes an inductor and first,second, third, fourth and fifth output nodes, the method comprising:configuring a switching network so that the boost converter operates ina first mode where the positive electrode of the inductor is connectedto an input voltage and the negative electrode of the inductor isconnected to ground; configuring the switching network so that the boostconverter operates in a second mode the negative electrode of theinductor is connected to ground and the positive electrode of theinductor is connected in sequence to one or more of the fourth and fifthoutput nodes; and configuring the switching network so that the boostconverter operates in a third mode where the positive electrode of theinductor is connected to the input voltage and the negative electrode ofthe inductor is connected in sequence to one or more of the first,second and third output nodes.
 20. A method as recited in claim 19 inwhich the first, second and third modes are selected in a repeatingsequence.
 21. A method as recited in claim 20 in which the repeatingsequence has the form: first mode, second mode, first mode, third mode.22. A method as recited in claim 20 in which the repeating sequence hasthe form: first mode, second mode, third mode.
 23. A method as recitedin claim 19 that further comprises modulating the duration of therespective connections between the inductor, the input voltage, groundand the output nodes to control the voltage of the output nodes.
 24. Amethod as recited in claim 19 that further comprises configuring theswitching network so that the boost converter operates in a fourth modewhere the negative electrode of the inductor is connected to ground andthe positive electrode of the inductor is connected in sequence to oneor more of the fourth and fifth output nodes.